The invention concerns a magnetic resonance (MR) detection configuration comprising at least one RF resonant circuit with an inductance, a preamplifier module, and an RF receiver, wherein a reactive transformation circuit is connected between a high-ohmic point of the inductance of the RF resonant circuit and a low-ohmic connecting point of the RF resonant circuit to act as an impedance transformer, and wherein the low-ohmic connecting point of the RF resonant circuit is connected to the preamplifier module via an RF line having a characteristic impedance.
A configuration of this type is disclosed in [4].
A sensor of an NMR spectrometer consists essentially of one radio frequency (RF) resonant circuit which is disposed closely around an NMR test sample. The RF resonant circuit normally functions as a transmitting antenna in a first time period, the so-called transmitting process. A strong pulse-shaped RF current is supplied to the RF resonant circuit, which generates a strong RF field in the NMR test sample and excites the nuclear spins contained therein. In a subsequent second time period, the so-called receiving process, the RF resonant circuit functions as a detector and receives the RF field radiated by the nuclear spins, the so-called FID signal (FID=free induction decay) which is supplied to the preamplifier and its matching network via an RF line.
In order to optimize the signal-to-noise ratio (=SINO) of the nuclear magnetic resonance signal (NMR signal), noise matching of the preamplifier to the RF line must be obtained. One end of the RF line which extends to the preamplifier may thereby not be terminated with the characteristic impedance of the RF line, which could cause standing waves to be generated in the RF line [4] during the receiving process. These influence the Q value and the resonance frequency of the loaded RF resonant circuit, such that the RF line and the matching network of the preamplifier must also be regarded as parts of the RF resonant circuit.
In order to obtain a high signal sensitivity, i.e. a high SINO of the FID signal, during the receiving process, the series loss resistance of the inductance of the RF resonant circuit must i.a. be minimum in order to also minimize the thermal noise produced therein. For this reason, cryogenically cooled RF resonant circuits of normally conducting materials and also of high temperature super conductor (HTSC) materials are used. This substantially reduces the noise and at the same time considerably increases the Q value of the unloaded RF resonant circuit (resonant circuit without cable and matching network).
However, RF resonant circuits with high Q values are disadvantageous in that, during the transmitting process, the pulse-shaped transmission energy cannot be immediately received by the RF resonant circuit or be removed therefrom, since the RF resonant circuit requires both a certain build-up time as well as decay time, which are longer the higher the Q value. A power-matched resonant circuit has twice the damping factor or half the time constant compared to a freely oscillating resonant circuit. This is easy to demonstrate. This double damping usually determines the build-up and decay behavior of the resonant circuit. This is, however, not sufficient e.g. for NMR spectroscopy applications.
The decay process is mostly critical, since it usually covers the start of the receiving process and thereby also the start of the FID signal, thereby preventing exact temporal separation between the transmission and receiving processes. For this reason, a third time period is suitably defined between the transmission and receiving process (called damping process below).
Additional complications result from the fact that modern NMR magnet systems have superconducting coils. For this reason, the space within the room temperature bore (RT bore), i.e. within the measuring space available, is very confined. In consequence thereof, the RF resonant circuit must be spatially separated from the preamplifier and be electrically connected to an RF line. Due to the high NMR frequencies of today's spectrometers, the length of this RF line may be a multiple of the wavelength of the NMR frequency. Depending on the construction and the NMR frequency, the RF line may, however, also be very short compared to the wavelength.
The problems, configurations and properties that occur in the three defined time periods are described below. In particular, the problems associated with RF resonant circuits having very high Q values are discussed.
During transmission, the width of the NMR spectrum that can be excited is limited by the available bandwidth of the transmission system and thereby, in particular, by the bandwidth of the RF resonant circuit used for transmission. A very high Q value of the RF resonant circuit may cause the bandwidth of the RF resonant circuit to be too small to excite the desired frequency range of the magnetic spins. Means must be found to nevertheless obtain the required excitation bandwidths of the NMR spectrum.
During damping, additional measures must be taken in order to maximally reduce the exponentially decaying process of the coil current in response to the transmission pulse, such that reception of the FID signal can be started as quickly as possible. This is very important, since the start of the FID signal contains i.a. very important information about the initial phase of the individual resonance frequencies which are contained in the FID signal. When this information is lost, the base line and the shape of the NMR resonance lines will be distorted in the NMR spectrum obtained after Fourier transformation (FT), and moreover the integral over the individual NMR lines can no longer be correctly calculated.
In addition to distortions in the spectrum, a long decay process also produces an undesired SINO loss, since no FID signal can be acquired during the decay process, i.e. part of the FID signal is lost.
The extent to which the decay process must have vanished before the receiving process can be started depends mainly on the preamplifier and the subsequent detection circuit. These must work in the linear region of their characteristic dependencies, i.e. must not be saturated. Acquisition of the FID signal may not be started before this condition is met.
The receiving process is a very complex process that substantially involves three problems:
1. The problem of radiation damping. This occurs when the spin concentration of the solvent or the test substance is very high. The integrating effect of the numerous magnetic spins in the test sample may be compared with a resonant circuit which is strongly electromagnetically coupled to the RF resonant circuit. In this case, the RF resonant circuit may react to this “resonant circuit”, which can cause broadenings, distortions and phase errors of the spectral lines.
When two resonant circuits are provided which are disposed at a certain separation from each other, their electromagnetic coupling and thereby their mutual influence increases, the larger the Q values of the individual resonant circuits. The situation for radiation damping is similar, with one resonant circuit being defined by the magnetic spins and the other by the RF resonant circuit. The first resonant circuit has a very high intrinsic effective Q value due to the properties of the magnetic spins. This Q value may be between 106 and 109. For this reason, the height of the Q value of the second resonant circuit, i.e. the RF resonant circuit, is important for observation of the coupling. This view is equivalent to the observation that, with a high Q value of the RF resonant circuit, large currents flow therein in response to the NMR signals, which again acts on the spins depending on the phase of the resonator current. In order to minimize radiation damping, this Q value should be minimum without thereby deteriorating the SINO.
2. The decay signal of the resonator in response to the transmission pulse is greatly reduced after damping but still has a residual portion that can cause distortions of the base line in the NMR spectrum. For this reason, it is important to also damp the RF resonant circuit during the receiving process. The time constant of the decay signal during the receiving process is thereby called τEV. The damping value that can be obtained during the receiving process is naturally smaller (i.e. the time constant is longer) than that during the damping process, since the circuit must be optimized primarily to an optimum SINO and not to optimum damping.
3. The third problem is caused when the Q value of the RF resonant circuit during the receiving process is too high for receiving the entire width of the desired NMR spectrum. Precise damping of the RF resonant circuit during the receiving process provides adjustment of the spectral receiving range of the RF resonant circuit to the width of the desired NMR spectrum. Such damping during the receiving process is only reasonable when the RF resonant circuit is also additionally damped during the transmission process, such that it yields the desired excitation bandwidth. The latter should be larger or equal to the width of the desired NMR spectrum.
It is important that damping of the RF resonant circuit does not deteriorate the SINO during the receiving process, which may seem contradictory at first, but can be addressed using so-called “electronic damping”. Damping of the RF resonant circuit is obtained by the electronic input impedance of the preamplifier [2]-[4]. The adjustment network of the preamplifier itself should, however, be noise-matched, i.e. the preamplifier itself should see optimum source impedance (the optimum source impedance is that source impedance that produces the highest SINO) downstream of its adjustment circuit.
In [1], the Q value of an RF resonant circuit 100′ is kept small during the transmission and damping process using a so-called “Q switch” (FIGS. 11a, 11b). The name already indicates that a resistance RQ is added to the RF resonant circuit 100′ using a switch 11. The resistance RQ and the switch 11 are connected in series and connected between the high-impedance point M of inductance L of the RF resonant circuit 100′ and ground. The resistance RQ damps the RF resonant circuit 100′, thereby reducing its Q value. The switch 11 is realized in most cases using a PIN diode DQ (FIG. 11b). However, switches with field effect transistors (FET) are also feasible. The PIN diode DQ is blocked with a high voltage HV which is applied in the reverse direction, or brought into a conducting state using a current I11BIAS in the forward direction.
This configuration is disadvantageous in that the additional components (resistance RQ, PIN diode DQ etc.) must be housed in the vicinity of the sensitive RF resonant circuit and can thereby deteriorate the homogeneity of the static magnetic field due to their magnetic susceptibility. Additionally, a control signal must be guided in the very sensitive region of the RF resonant circuit. Additional wiring of the RF resonant circuit unavoidably reduces the Q value during the receiving process and thereby produces an undesired SINO loss. The parallel loss resistance of the PIN diode DQ in the blocked state is sufficient to cause this loss.
Moreover, the PIN diode DQ of cryogenically cooled RF resonant circuits should also be cooled. This is possible only to a limited degree due to the “carrier-freeze-out-effect” in semiconductors, which is very problematic.
The Q switch cannot be directly used for damping the RF resonant circuit during the receiving process, since it would cause an excessive SINO loss. Special methods have been developed with which the Q switch is intermittently used, i.e. is switched on temporarily within a time interval between two data points and subsequently switched off again prior to detection of the next data point. This method also produces a SINO loss, since there is less time for data acquisition.
The configuration used in [2] consists of an RF resonant circuit and a preamplifier which are located close to each other and therefore need not be connected to each other via an RF line. The preamplifier is negative feedbacked and thereby generates an input impedance which depends on the size of the feedback. This input impedance is used as a damping impedance for the RF resonant circuit during the receiving process and is directly connected to the high-impedance point of the RF resonant circuit. Since the damping impedance is generated electronically by active elements, this type of damping is also called “electronic damping”. In the conventional circuit, two separate couplings to the RF resonant circuit are used: the first for the transmission process and the second for the damping and receiving processes. Both couplings are provided at the high-impedance point of the RF resonant circuit. During the transmission process, the RF resonant circuit is damped by normal power matching.
During the decay time of the RF resonant circuit (damping process), the latter practically remains undamped throughout the entire time, since the electronic damping effect is cancelled by limiting diodes for safety reasons, as is explained below. It is only at the very end of the decay time, that the electronic damping again becomes effective, but can no longer cause any significant reduction of the overall time of the decay process.
The preamplifier is optimally noise-matched during the receiving process. When the quotient between the magnitude of the input impedance and the magnitude of optimum source impedance of the preamplifier differs greatly from 1, the RF resonant circuit can also be damped without any SINO loss. The RF resonant circuit is additionally loaded by the two adjustment paths during the transmission and receiving process, which nevertheless finally produces a SINO loss, mainly when the Q values of the RF resonant circuit are very high.
Today's preamplifiers usually require limiting diodes at the input, which should protect against overloading during the transmission process. These act more or less like low-ohmic voltage sources of approximately 2 V and thereby prevent the occurrence of excessive voltages at the sensitive input of the preamplifier during the transmission pulse and the subsequent decay process. In consequence thereof, the electronic damping by the input impedance of the preamplifier becomes active and provides its damping effect only when the preamplifier with limiting diodes is operating in a linear range. The main portion of the decay process is therefore not additionally damped in the present configuration, which operates only with an electronic impedance as damping means, and the coil energy dies down with the undamped time constant.
Moreover, in current NMR spectrometers comprising high field superconducting magnets, it is practically not possible to closely arrange the RF resonant circuit and preamplifier, since there is no space for the preamplifier in the direct vicinity of the RF resonant circuit. The magnetic field strength of the NMR magnet is also too high for unproblematic function of the preamplifier and moreover cryogenically cooled RF resonant circuits and preamplifiers require different cryogenic temperatures for optimum operation, i.e. both units would have to be thermally insulated from each other, which, in turn, would cause space problems.
Additionally, the above-described feedback of the preamplifier practically cannot be realized in today's high field systems for reasons of stability. The associated high NMR frequencies e.g. in a 900 MHz NMR spectrometer the feedback of the preamplifier may produce a strong oscillation tendency which can be eliminated only with great effort, if at all.
For all above-mentioned reasons, the configuration disclosed in [2] is not suitable for today's high field NMR spectrometers.
[3] Discloses a configuration that comprises an RF resonant circuit and a preamplifier which are also disposed close to each other. It mentions the possibility of spatial separation of both objects and connecting them via an RF line, but does not mention the associated problems (treatment of optimum damping under the limitation that results from adjustment to the line of impedance RW). As in the configuration of [2], the configuration of [3] also achieves electronic damping of the RF resonant circuit using the input impedance of the active part of the preamplifier (without matching network), wherein this damping causes no loss in optimum SINO and is stronger the more the ratio between the input impedance and the optimum noise impedance of the preamplifier differs from 1 (overcoupling). The RF resonant circuit is coupled at the low-impedance point A of the RF resonant circuit. Different types of impedance transformation between point A and the high-impedance point M of the RF resonant circuit are proposed and examined in view of the SINO.
In the configuration of [3], the decay time is reduced by 30%. This is, however, far too little for today's high field systems. Rather, reductions of approximately 90% are desired. Moreover, with large “overcoupling”, undesired effects occur, e.g. saturation effects in the preamplifier and the decay times can increase again. The decay process may even contain two frequency components whose envelope decays very slowly.
The configuration disclosed in [4] also comprises one RF resonant circuit 100 and a preamplifier module 2 which are, however, positioned at a separation from each other and are connected via an RF line 15 (FIG. 10). The RF resonant circuit 100 comprises an inductance L and a loss resistance RS and is connected at a low-impedance connecting point A to a preamplifier module 2 via the RF line 15 of line impedance RW, which comprises a preamplifier 5 with an matching network AN and with the active part 5′ of the preamplifier.
In order to maintain the SINO, the impedance transformation of the loss resistance RS of the inductance L of the RF resonant circuit 100 relative to a point V between the matching network AN and the input of the active part 5′ of the preamplifier 5 must produce an impedance value which is equal to the optimum source impedance of the active part of the preamplifier 5′. When this condition is met, and when the ratio between the magnitudes of the optimum source impedance and the input impedance of the active part of the preamplifier 5′ differs greatly from 1, the transform of the input impedance of the preamplifier 5 relative to the low-impedance connecting point A of the RF resonant circuit 100 differs greatly from the line impedance RW of the RF line 15 between the RF resonant circuit 100 and the preamplifier module 2 and thereby provides much stronger damping of the RF resonant circuit 100 compared to conventional power matching.
The impedance transformation between the RF resonant circuit 100 and the preamplifier 5 is performed such that, in a first step, the loss resistance RS of the inductance L of the RF resonant circuit 100 is transformed to a value RA at the low ohmic connecting point A using a coupling capacitor. RA is usually equal to RW. The RF line 15 which has the impedance RW is thereby matched at the low ohmic connecting point A without reflection such that the loss in the RF line is reduced.
When viewing from point B, disposed at the end of the RF line 15 facing away from the RF resonant circuit, towards the line 15, the resistance value RW appears. It is transformed in a second step by means of the matching network AN such that the active part of the preamplifier 5′ sees the optimum source impedance at point V.
[4] thus describes extensive measures for reducing radiation damping during the receiving process. These measures may also have a strong damping effect on the RF resonant circuit during the receiving process without producing a SINO loss. Coupling to the RF resonant circuit is effected at the low impedance connecting point A of the RF resonant circuit. Optimum damping of the RF resonant circuit during the receiving process and the produced advantages for decay of the RF resonant circuit is, however, not mentioned.
The Q value of the RF resonant circuit plays a central role in the design of a transmitting/receiving system for NMR and for this reason there is great need for a method that optimally reduces the disturbing influences of the high Q value of the RF resonant circuit during the transmission, damping and receiving process, wherein the SINO must not decrease during the receiving process.
The above-described prior art only provides solutions with electronic damping impedances (input impedances of preamplifiers) for the damping process and when damping is performed at the low impedance point A of the RF resonant circuit, which do not optimally damp the RF resonant circuit due to their non-linear properties, since damping acts, in particular, only when the amplitude of the oscillation has become sufficiently small that it is in the linear range of the preamplifier. Towards this end, it must have largely decayed already. Thus, most of the time passes without damping.
The conventional proposed solutions merely present individual solutions which are either optimized for the transmission and damping process or for the receiving process, but do not provide a comprehensive solution that provides optimum relationships in all three processes.
It is therefore the underlying purpose of the invention to provide an MR detection configuration, wherein the RF resonant circuit and the preamplifier module are spatially separated from each other, comprising an extensive damping concept, wherein all three processes (transmitting, damping and receiving process) are optimized.